Deeply integrated voltage regulator architectures

ABSTRACT

A system is disclosed. The system includes a substrate, and a first chip on the substrate, where a load circuit is integrated on the first chip. The system also includes a second chip on the substrate, where a power delivery circuit is configured to deliver current to the load circuit according to a regulated voltage at a node. The power delivery circuit includes a first circuit configured to generate an error signal based at least in part on the regulated voltage, and a voltage generator including power switches configured to modify the regulated voltage according to the error signal, where the first circuit of the power delivery circuit is integrated on the first chip, and where at least a portion of the power switches of the power delivery circuit are integrated on the second chip.

CROSS-REFERENCES TO RELATED APPLICATIONS

This application claims the benefit of U.S. Provisional application No.62/785,143, filed Dec. 26, 2018, titled “INTEGRATED VOLTAGE REGULATOR,”the disclosure of which is incorporated herein by reference.

FIELD OF THE INVENTION

The present application generally pertains to power delivery circuits,and more particularly to circuits which deliver power to a load usingmultiple phases.

BACKGROUND OF THE INVENTION

New circuits have increased power needs. Therefore, power deliverysystems having improved control schemes are needed.

BRIEF SUMMARY OF THE INVENTION

One inventive aspect is a system. The system includes a substrate, and afirst chip on the substrate, where a load circuit is integrated on thefirst chip. The system also includes a second chip on the substrate,where a power delivery circuit is configured to deliver current to theload circuit according to a regulated voltage at a node. The powerdelivery circuit includes a first circuit configured to generate anerror signal based at least in part on the regulated voltage, and avoltage generator including power switches configured to modify theregulated voltage according to the error signal, where the first circuitof the power delivery circuit is integrated on the first chip, and whereat least a portion of the power switches of the power delivery circuitare integrated on the second chip.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 depicts an illustrative simplified schematic of a system.

FIG. 2 is a cross-sectional schematic view of a IC package.

FIG. 3 depicts an illustrative simplified schematic of a power deliverycontrol circuit that can be used in a variety of electronic systems.

FIG. 4 is a waveform diagram illustrating wave forms for signals of thepower delivery control circuit 100 illustrated in FIG. 1.

FIG. 5 is a waveform diagram illustrating wave forms for signals of thepower delivery control circuit 100 illustrated in FIG. 1.

FIG. 6 is a diagram illustrating Tc (time between starts of phasepulses) and Verr (error voltage) dependence on load current.

FIG. 7A is a schematic illustration of a power delivery engine.

FIG. 7B illustrates one example of the waveforms for the power deliveryengine 500 illustrated in FIG. 7A.

FIG. 8 is a schematic illustration of a power delivery engine.

FIG. 9 is a schematic illustration of a control timer circuit.

FIG. 10 is a schematic illustration of a comparator mode controlcircuit.

FIGS. 11 and 12 illustrate an embodiment of a voltage to time circuit.

FIGS. 13 and 14 illustrate an embodiment of a voltage to time circuit.

FIGS. 15 and 16 illustrate an embodiment of inductor shorting.

FIG. 17 is a schematic illustration of a control timer circuit.

FIGS. 18-20 are schematic illustrations of compensation networksaccording to some embodiments.

FIG. 21 is a flowchart of a repetitive switching sequence providing acontinuous current output for the switched regulation circuit in FIG. 5Aaccording to an embodiment of the invention.

FIG. 22 is a timing diagram of voltages and currents within the switchedregulation circuit of FIG. 7A according to the switching sequence inFIG. 21.

FIG. 23 is a schematic of the switched regulation circuit shown in FIG.7A in a particular switch configuration according to the switchingsequence in FIG. 21.

FIG. 24 is a schematic of the switched regulation circuit shown in FIG.7A in a particular switch configuration according to the switchingsequence in FIG. 21.

FIG. 25 is a schematic of the switched regulation circuit shown in FIG.7A in a particular switch configuration according to the switchingsequence in FIG. 21.

FIG. 26 is a schematic of the switched regulation circuit shown in FIG.7A in a particular switch configuration according to the switchingsequence in FIG. 21.

FIG. 27 is a schematic of the switched regulation circuit shown in FIG.7A in a particular switch configuration according to the switchingsequence in FIG. 21;

FIG. 28 is a schematic of the switched regulation circuit shown in FIG.7A in a particular switch configuration according to the switchingsequence in FIG. 21.

DETAILED DESCRIPTION

Particular embodiments of the invention are illustrated herein inconjunction with the drawings.

FIG. 1 depicts an illustrative simplified schematic of system 10including a power delivery circuit 20 that is coupled to a load 12,which receives power from power delivery circuit 20. For example, load12 may receive current from power delivery circuit 20 via node Vout,where the voltage at node Vout is regulated by power delivery circuit20. Accordingly, as understood by those skilled in the art, load 12 is aload for power delivery circuit 20.

The load 12 is integrated on a first semiconductor chip or die. A firstportion of the power delivery circuit 20 is also integrated on the firstsemiconductor chip or die. In addition, a second portion of the powerdelivery circuit 20 is integrated on a second semiconductor chip or die.

In some embodiments, as described above, the load 12 can be any type ofintegrated circuit (e.g., a processor, DSP, AI computation,communication) and may contain digital and analog circuits. The load 12may interface with the power delivery circuit 20. The means ofcommunication can include a communication bus coupled to either or bothof the first and second portions of power delivery circuit 20. Forexample, the load 12 can include any portion of a monitoring or controlsystem that interfaces with one or more portions of the power deliverycircuit 20. The load 12 can also store information associated with thepower delivery circuit 20 in a memory and communicate data associatedwith the power delivery circuit 20 to an external device.

The load 12, can include circuitry that configures the power deliverycircuit 20. For example in can communicate the desired output voltage.It may also configure properties of the power delivery circuit 20including compensation information, startup and shutdown informationetc. The load can communicate to the power delivery circuit on eitherthe second die or the first die. A dedicated communication connectionmay also be connected between parts of the power delivery circuit 20 onthe first and second die.

Communication between circuits on the same die can often be much faster.Being able to change the voltage quickly can increase throughput andsave power so one implementation enables the load to quickly change thevoltage desired by communicating with the portion of the power deliverycircuit on the first die. In some implementations, the desired voltagemay also be communicated to the portion of the power delivery circuit 20on the second die. This can enable optimal optimization of thatcircuitry. Other information that can be communicated to the powerdelivery circuit 20 include power states of the load, timing and rate ofstartup and shut down of the power delivery circuits. Other informationthat can be communicated from the power delivery circuit 20, to the load12 include status of the regulator, information about the outputvoltage, how much current is being provided, temperature and any faultconditions. This information may be used by the receiving circuit orcomponent to modify the functionality of the receiving component. Forexample, the information may be used by the receiving circuit orcomponent to modify the voltage generated by power delivery circuit 20.

The power delivery circuit 20 includes error circuit 22, errormanagement circuit 24, switch control circuit 26, power switches 28,inductors 38, and capacitor 32. In some embodiments, the power deliverycircuit 20 forms or partly forms a voltage regulator circuit.

Error circuit 22 is configured to receive a reference voltage at node orbus Vref. The reference voltage is generated by another circuit, and hasa voltage value equal or substantially equal to a target or desiredvoltage value of the voltage at node Vout, which is generated by powerdelivery circuit 20 for load 12, such that the power delivery circuit 20delivers current to the load 12 at a regulated voltage at node Vout,where the regulated voltage at node Vout is determined based on thereference voltage at node or bus Vref.

Error circuit 22 is configured to also receive the voltage at node Voutgenerated by power delivery circuit 20. Based at least on the differencebetween the voltage at node Vout and the reference voltage, errorcircuit 22 generates an error signal which causes the power deliverycircuit 20 to generate the voltage at node Vout so that the differencebetween the voltage at node Vout and the reference voltage is minimized,as understood by those of skill in the art.

In some embodiments, error circuit 22 comprises an operational amplifierhaving gain, bandwidth, and stability characteristics which contributeto stable generation of the voltage at node Vout.

Error circuit 22 may include an analog-to-digital converter configuredto generate a digital representation of the difference between theanalog voltage at node Vout and the analog reference voltage at nodeVref as the error signal.

In some embodiments, error circuit 22 comprises an analog-to-digitalconverter configured to generate a digital representation of the voltageat node Vout, and a digital difference or subtraction circuit configuredto receive the digital representation of the voltage at node Vout and togenerate a digital error signal based on the difference between thedigital representation of the voltage at node Vout and the referencevoltage, as represented by a digital word received by error circuit 22at node or bus Vref.

Alternatively, in some embodiments, error circuit 22 comprises an analogamplifier configured to receive the analog voltage at node Vout and ananalog reference voltage at node Vref, and to generate an analog errorsignal. In some implementations, the analog error signal represents thedifference between the analog voltage at node Vout and the analogreference voltage at node Vref. In some implementations, the analogerror signal represents the difference between the analog voltage atnode Vout and the analog reference voltage at node Vref multiplied by again factor. In some implementations the amplifier can includecompensation as known in the art. For example the amplifier can beconfigured to have proportional, integral and/or differential processingof the analog voltage at node Vout. In other implementations the errorcircuit 22 can be, or include with other circuits, a comparator togenerate an error signal indicating whether the analog voltage at nodeVout is greater than or less than a reference voltage, such as thereference voltage at node Vref.

At least a portion of error circuit 22 may be integrated on the firstsemiconductor chip or die, whereon the load 12 or at least a portion ofthe load 12 is also integrated. In some embodiments, all of errorcircuit 22 is integrated on the first semiconductor chip or die. In someembodiments, at least a portion of error circuit 22 is integrated on thesecond semiconductor chip or die. In some embodiments, at least aportion of error circuit 22 is integrated on the second semiconductorchip or die with one or more other portions of the power deliverycircuit 20.

It may be advantageous to integrate at least a portion of error circuit22 on the first semiconductor chip or die at least because error circuit22 or at least a portion of error circuit 22 comprises a sensorconfigured to sense the voltage at node Vout for power delivery circuit20. As understood by those of skill in the art, power delivery circuit20 regulates the voltage at node Vout based on the difference betweenthe voltage at node Vout and the voltage reference voltage at node Vref.Accordingly, it is advantageous for the sensor to be as close aspossible to the optimum point of sensing the voltage at node Vout, whichis at the load, as understood by those of skill in the art, so that thesensed value is as accurate as possible. As understood by those of skillin the art, distance between the optimum point and the sensed point atnode Vout allows for the sensed voltage to be different from the actualvoltage at least because of, for example, noise and IR drop.

Error management circuit 24 is configured to receive the output(s) oferror circuit 22 and to generate one or more signals based on the outputof error circuit 22. The one or more signals generated by errormanagement circuit 24 influence the voltage at node Vout so as tominimize the difference between the voltage at node Vout and thereference voltage, as understood by those of skill in the art.

In some embodiments, the output of error management circuit 24 is arepresentation of the difference between the voltage at node Vout andthe reference voltage.

The output of error management circuit 24 may include one or more of ananalog voltage, a digital word, and another signal type. For example, insome embodiments, the output of error management circuit 24 may includea series of digital pulses, where the frequency of the pulsescorresponds at least with the relationship (e.g. difference) between thevoltage at node Vout and the reference voltage. In alternativeembodiments, other signal mechanisms, for example encoding thedifference between the voltage at node Vout and the reference voltage,are generated by error management circuit 24.

In some embodiments, the error management circuit 24 receives arepresentation of the current being provided to the load, for example,from the load 12, the error circuit 22, or switch control circuit 26,and the output of the error management circuit 24 is determined based onthe current being drawn from or supplied to the load and arepresentation of the voltage at node Vout.

At least a portion of error management circuit 24 may be integrated onthe first semiconductor chip or die, whereon the load 12 or at least aportion of the load 12 is also integrated. In some embodiments, all oferror management circuit 24 is integrated on the first semiconductorchip or die. In some embodiments, at least a portion of error managementcircuit 24 is integrated on the second semiconductor chip or die. Insome embodiments, at least a portion of error management circuit 24 isintegrated on the second semiconductor chip or die with one or moreother portions of the power delivery circuit 20. In some embodiments,all of error management circuit 24 is integrated on the secondsemiconductor chip or die.

Switch control circuit 26 is configured to receive the output of errormanagement circuit 24 and to generate one or more signals based on thereceived output of error management circuit 24. The one or more outputsignals generated by the switch control circuit 26, for example, withpower FET drivers, influences the voltage at node Vout so as to minimizethe difference between the voltage at node Vout and the referencevoltage, as understood by those of skill in the art.

In some embodiments, the output of switch control circuit 26 is based onthe difference between the voltage at node Vout and the referencevoltage. In some embodiments, the output of switch control circuit 26 isbased additionally or alternatively on a current delivered to load 12.

The output of switch control circuit 26 may include signals from powerFET drivers which control the conductivity states of the power switches28. For example, in some embodiments, the output of switch controlcircuit 26 may include multiple series of digital pulses, where thefrequency and the timing relationship among the multiple seriescorresponds with at least the difference between the voltage at nodeVout and the reference voltage. In alternative embodiments, other signalmechanisms, for example corresponding with the difference between thevoltage at node Vout and the reference voltage, are generated by switchcontrol circuit 26.

In some embodiments, the switch control circuit 26 includes currentsense circuitry, which senses the current through power switches 28. Asunderstood by those of skill in the art, the sensed current may be usedto influence when the power switches 28 should switch.

In some embodiments, switch control circuit 26 generates signalsrepresenting the current being delivered to the load for either or bothof error circuit 22 and error management circuit 24.

At least a portion of switch control circuit 26 may be integrated on thefirst semiconductor chip or die, whereon the load 12 or at least aportion of the load 12 is also integrated. In some embodiments, all ofswitch control circuit 26 is integrated on the first semiconductor chipor die. In some embodiments, at least a portion of switch controlcircuit 26 is integrated on the second semiconductor chip or die. Insome embodiments, at least a portion of switch control circuit 26 isintegrated on the second semiconductor chip or die with one or moreother portions of the power delivery circuit 20. In some embodiments,all of switch control circuit 26 is integrated on the secondsemiconductor chip or die.

Power switches 28 are configured to receive the output of switch controlcircuit 26 and to cooperatively generate the voltage at node Vout withinductor(s) 30 and capacitor 32, wherein the generated voltage is basedon the received output of switch control circuit 26. Accordingly, powerswitches 28, inductor(s) 30, and capacitor 32 collectively form avoltage generator driven and controlled by the output of switch controlcircuit 26. Therefore, the output of switch control circuit 26 causesthe power switches 28, inductor(s) 30, and capacitor 32 to influence thevoltage at node Vout so as to minimize the difference between thevoltage at node Vout and the reference voltage, as understood by thoseof skill in the art.

In some embodiments, power switches 28 are configured to selectively,alternately, and repetitively connect inductor(s) 30 to a positive powersupply and to a negative power supply. In some embodiments, switchcontrol circuitry 26, power switches 28, inductor(s) 30, and capacitor32 are arranged so as to form a synchronous buck converter topology, afull-bridge converter topology, a boost topology, a buck-boost oranother type of power regulator circuit topology.

Power switches 28 may be integrated on the second semiconductor chip ordie. In some embodiments, power switches 28 are discrete power devicesor are integrated on another semiconductor die.

Inductors 38 and capacitor 32 collectively form an LC filter, asunderstood by those of skill in the art. In some embodiments, capacitor32 is implemented as multiple physical capacitors connected in parallel.Inductors 38 and capacitor 32 or portions of inductors 38 and capacitor32 may be formed on either of the first and second semiconductor chipsor die. In some embodiments, the output capacitor 32 is only located onthe first semiconductor die. In some embodiments, the output capacitor32 is a combination of capacitors some on the first semiconductor die,with some outside of the first semiconductor die. In some embodiments,either or both of inductors 38 and capacitor 32 are placed within amulti-chip or multi-die package carrying the first and secondsemiconductor chips or die, and are not integrated on either of thefirst and second semiconductor chips or die. In some embodiments, eitheror both of inductors 38 and capacitor 32 are placed outside of amulti-chip or multi-die package carrying the first and secondsemiconductor chips or die.

In some embodiments, power delivery system 20 is multi-phase. In suchembodiments, inductor(s) 30 comprises multiple inductors, each connectedto capacitor 32 and to a distinct pair of power switches of powerswitches 28. Each distinct pair of power switches of power switches 28is separately controlled by switch control circuitry 26 such thatmulti-phase power is delivered by power switches 28, inductor(s) 30, andcapacitor 32 to load 12.

In some embodiments, error management circuit 24 and switch controlcircuit 26 are integrated into a loop control circuit having the inputfunctionality of error management circuit 24 and the outputfunctionality of switch control circuit 26.

In such embodiments, at least a portion of the loop control circuit maybe integrated on the first semiconductor chip or die, whereon the load12 or at least a portion of the load 12 is also integrated. In someembodiments, all of the loop control circuit is integrated on the firstsemiconductor chip or die. In some embodiments, at least a portion ofthe loop control circuit is integrated on the second semiconductor chipor die. In some embodiments, at least a portion of the loop controlcircuit is integrated on the second semiconductor chip or die with oneor more other portions of the power delivery circuit 20.

As discussed above, at least in some embodiments, error circuit 22receives analog voltages at nodes Vout and Vref, and switch controlcircuit 26 generates control signals at for power switches 28.Accordingly, error circuit 22, error management circuit 24, and switchcontrol circuit 26 may be considered as collectively generating thecontrol signals. Accordingly, error circuit 22, error management circuit24, and switch control circuit 26 may be collectively considered acontrol signal generator.

In such embodiments, at least a portion of the collective control signalgenerator control signal generator may be integrated on the firstsemiconductor chip or die, whereon the load 12 or at least a portion ofthe load 12 is also integrated. In some embodiments, all of thecollective control signal generator is integrated on the firstsemiconductor chip or die. In some embodiments, at least a portion ofthe collective control signal generator is integrated on the secondsemiconductor chip or die. In some embodiments, at least a portion ofthe collective control signal generator is integrated on the secondsemiconductor chip or die with one or more other portions of the powerdelivery circuit 20.

As understood by those of skill in the art, the collective controlsignal generator converter may be segregated into an analog circuitportion and a digital circuit portion. At least a portion of the analogcircuit portion may be integrated on the first semiconductor chip ordie, whereon the load 12 or at least a portion of the load 12 is alsointegrated. In some embodiments, all of the analog circuit portion isintegrated on the first semiconductor chip or die, for example, suchthat a digital signal is sent from the first semiconductor chip or dieto the second semiconductor chip or die. In some embodiments, at least aportion of the analog circuit portion is integrated on the secondsemiconductor chip or die. In some embodiments, at least a portion ofthe analog circuit portion is integrated on the second semiconductorchip or die with one or more other portions of the power deliverycircuit 20.

In some embodiments, the signal from the first die to the second dieindicates when a phase should start a new pulse. The second die may takethis signal and decide which phase should switch next so as to minimizethe voltage ripple at the output voltage at the load 12.

In some embodiments, system 10 also includes a reference voltagegeneration circuit (not shown), which is configured to generate thereference voltage at node Vref.

In such embodiments, at least a portion of the reference voltagegeneration circuit may be integrated on the first semiconductor chip ordie, whereon the load 12 or at least a portion of the load 12 is alsointegrated. In some embodiments, all of the reference voltage generationcircuit is integrated on the first semiconductor chip or die. In someembodiments, at least a portion of the collective analog-to-digitalconverter is integrated on the second semiconductor chip or die. In someembodiments, at least a portion of the reference voltage generationcircuit is integrated on the second semiconductor chip or die with oneor more other portions of the power delivery circuit 20.

In some embodiments, current provided to load 12 may additionally oralternatively be used in any of error circuit 22, error managementcircuit 24, and switch control circuit 26 to generate the controlsignals for the power switches 28. Sensing the current provided to load12 can be done in either of the first and second die. The currentprovided to load 12 may also be sensed by sensing the voltage across theinductor(s) 30.

FIG. 2 is a cross-sectional schematic view of a IC package 50 havingpower delivery system 20 and load 12. As shown, IC package 50 includesfirst and second die 51 and 52 on substrate 55. As understood by thoseof skill in the art, first and second die 51 and 52 may be electricallyconnected to one another, for example, through connectors on the firstand second die and metal in the a package or printed circuit board.Additionally, first and second die 51 and 52 may be electricallyconnected to conductive pins or leads which extend so as to be exposedto components external to package 50 with one or more bond wires orother conductors. For example, first and second die 51 and 52 may beelectrically connected with a communication bus.

Load 12 or at least a portion of load 12 may be integrated on the firstdie 51. In addition, power switches 28 may be integrated on the seconddie 52.

In some embodiments, a first portion of error circuit 22 is integratedon the first die 51, where a second portion of error circuit 22 isintegrated on the second die 52 along with error management circuit 24and switch control circuit 26.

In some embodiments, error circuit 22 is entirely integrated on thefirst die 51, and error management circuit 24 and switch control circuit26 are integrated on the second die 52.

In some embodiments, a first portion of error management circuit 24 isintegrated on the first die 51 along with error circuit 22, where asecond portion of error management circuit 24 is integrated on thesecond die 52 along with switch control circuit 26.

In some embodiments, error circuit 22 and error management circuit 24are entirely integrated on the first die 51, and switch control circuit26 is integrated on the second die 52.

In some embodiments, a first portion of switch control circuit 26 isintegrated on the first die 51 along with error management 24 and errorcircuit 22, where a second portion of switch control circuit 26 isintegrated on the second die 52.

In some embodiments, switch control circuit 26, error circuit 22, anderror management circuit 24 are entirely integrated on the first die 51.

In some embodiments, the inductors 30 are integrated on the second die52.

In some embodiments, the inductors 30 are external to both the first andsecond die 51 and 52. For example, the inductors 30 may be formed by ametal on substrate 55 external to both the first and second die 51 and52, and electrically connected to either or both of the first and seconddie 51 and 52. Alternatively, the inductors 30 may be connected externalto package 50, and electrically connected to either or both of the firstand second die 51 and 52.

In some embodiments, the capacitor 32 is integrated on the second die52.

In some embodiments, the capacitor 32 is external to both the first andsecond die 51 and 52. For example, the capacitor 32 may be connected tosubstrate 55 external to both the first and second die 51 and 52, andelectrically connected to either or both of the first and second die 51and 52. Alternatively, the capacitor 32 may be connected external topackage 50, and electrically connected to first die 51 and/or to seconddie 52.

In some embodiments, the capacitor 32 is integrated solely on the seconddie 52.

In some embodiments power switches 28 are on integrated on the seconddie 52.

In some embodiments, for example, in embodiments using a boost regulatorconfiguration, the power switches 28 are partially integrated on thesecond die 52, and are partially integrated on the first die 51.

In some embodiments only the power switches of the power delivery system20 are integrated on the second die 52, and the remainder of the powerdelivery system 20 is integrated on the first die 51.

A separate fabrication process can be used to form the first and seconddie 51 and 52, where the fabrication process of the first die isoptimized for forming power semiconductors. This is one of theadvantages of the disclosed allocations of circuitry onto the first andsecond die 51 and 52. Circuitry that is preferentially fast can beintegrated on to the second die 52, where the process may be better forfast operation. The first die 51 can be in a process that is optimizedfor creating power switches.

In some embodiments, first and second die are positioned within 200-500microns of one another. In some embodiments, the first and second dieare close enough that the first and second die use high speedcommunications that enables the power delivery circuit 20 to deliverpower to the load 12 without a capacitor 32 external to the first andsecond die or without a capacitor 32 external to package 50. Forexample, capacitor 32 may be formed on the first die, on the second die,or within package 50 external to the first and second die. In someembodiments, capacitor 32 within the electronic package can be in theform of one or more discrete capacitors positioned within the electronicpackage.

In some embodiments the inductor that couples power to the load can bean air-core inductor, that is, an inductor without a magnetic core.Various embodiments can have a reduced need for capacitance of capacitor32, as compared to traditional architectures. Some embodiments havesubstantially reduced parasitics and delays, as compared to traditionalarchitectures, allowing the power delivery circuit 20 to respond muchfaster to transient power requirements of the load 12.

The load 12 and/or the portion of the power delivery circuit 20integrated on the first die can communicate with and control the portionof the power delivery circuit 20 integrated on the second die via acommunication bus. In some embodiments the communication bus is analogwhile in other embodiments it is digital, including but not limited toan I2C bus. In some configurations it is a high speed digital bus. Invarious embodiments the communication bus can be bi-directional suchthat the load 12 and/or the portion of the power delivery circuit 20integrated on the first die can send signals to the portion of the powerdelivery circuit 20 integrated on the second die, and the portion of thepower delivery circuit 20 integrated on the second die can send signalsto the load 12 and/or the portion of the power delivery circuit 20integrated on the first die. In some embodiments, the communication bushas one, two, three, four or more separate conductors. In someembodiments the communication bus has one or more of the followingarchitectures and/or features:

In some multiphase embodiments, the number of phases of the powerdelivery circuit 20 is not communicated to the load 12 and/or theportion of the power delivery circuit 20 integrated on the first die.

In some embodiments, the load 12 and/or the portion of the powerdelivery circuit 20 integrated on the first die sends a signal to theportion of the power delivery circuit 20 integrated on the second dieevery time a pulse should be started (e.g., PWM Signal, off to high sideon, or low side on to high side on, high side on to low side, or start aresonant pulse).

In some embodiments, the load 12 and/or the portion of the powerdelivery circuit 20 integrated on the first die sends a signal to theportion of the power delivery circuit 20 integrated on the second dieindicating which phase should be triggered next.

In some embodiments, the load 12 and/or the portion of the powerdelivery circuit 20 integrated on the first die sends a signal to theportion of the power delivery circuit 20 integrated on the second dieindicating information related to the output voltage. That informationcan contain one or more of: an error voltage, an absolute voltage, acurrent of the load, a digital representation of the output voltage, adigital representation of the error voltage, a processed (i.e.,compensated version) of the error voltage that is an output of thecompensation circuit, or any other information.

In some embodiments, the load 12 and/or the portion of the powerdelivery circuit 20 integrated on the first die sends a signal to orreceives a signal from the portion of the power delivery circuit 20integrated on the second die indicating the value of the referencevoltage at node Vref.

In some embodiments, the load 12 and/or the portion of the powerdelivery circuit 20 integrated on the first die sends a signal to theportion of the power delivery circuit 20 integrated on the second dieindicating status information (e.g., how much current is beingdelivered, temperature, etc.).

In some embodiments, the load 12 and/or the portion of the powerdelivery circuit 20 integrated on the first die sends a signal to theportion of the power delivery circuit 20 integrated on the second dieindicating configuration information (e.g., desired voltage,compensation settings, power saving settings).

In some embodiments, the load 12 and/or the portion of the powerdelivery circuit 20 integrated on the first die sends a signal to orreceives a signal from the portion of the power delivery circuit 20integrated on the second die with information including:

A desired power state of the portion of the power delivery circuit 20integrated on the first die (e.g. off, low power, high power, voltage ofthe integrated portion of the power delivery circuit).

On and off signals for each phase and/or each individual power switchintegrated on the second die.

The state of one or more portions of the power delivery circuit 20(e.g., current, temperature, voltage, current, error state, etc.).

FIG. 3 is a schematic illustration of a system including power deliverycircuit 100 and load 175. The system of FIG. 3 is an example embodimentof system 10 of FIG. 1. Other embodiments can be used.

FIG. 3 depicts an illustrative simplified schematic of a power deliverycontrol circuit 100 that can be used in a variety of electronic systems.As shown in FIG. 1, the power delivery control circuit includes threephases 110, 120, and 130, however in other embodiments the powerdelivery control circuit can have from one to any number of phases.Phases 110, 120, and 130 are collectively an embodiment of powerswitches 28 or power switches 28 and switch control 26 of system 10. Insome embodiments each phase can include one or more solid-state switchesthat regulate power delivered from a power source to a load. In variousembodiments each phase can include an arrangement of sequentiallycoupled solid-state switches while in further embodiments each phase caninclude a pair of solid-state switches arranged in a synchronous buckconverter topology, while in yet further embodiments each phase can be afull-bridge or other type of power regulator circuit.

As further illustrated in FIG. 3, each phase is coupled to andcontrolled by a control circuit 150. The control circuit 150 includesVout and Vref nodes as inputs into an error amplifier 152. Erroramplifier 152 is an embodiment of error circuit 22 of system 10. Theerror amplifier 152 generates an error voltage at node Verr based on thedifference between Vout and Vref inputs. The error voltage can be usedas an input into a Voltage to Time circuit 154, which is an embodimentof error management circuit 24 of system 10. In some embodiments,instead of using the error voltage as an input to the Voltage to Timecircuit 154, a signal that is derived from the error voltage can be usedas an input. In one example the signal can be derived from a Type 2compensation network. The Voltage to Time circuit 154 is configured toconvert the error voltage, or the signal derived from the error voltage,to a controlled time (Tc). In some embodiments, the Voltage to Timecircuit 154 sends a series of clock pulses to a phaser circuit 156,where the time between the beginning of the pulses is equal orsubstantially equal to the controlled time (Tc). Phase circuit 156 is anembodiment of switch control circuit 26.

When the phaser circuit 156 receives each clock pulse signal, itdetermines which of phases 110, 120, and 130 to trigger next and sends atrigger signal to the determined phase. For example, at very light loadsonly phase 110 may be repetitively triggered so the phaser circuit 156only sends trigger voltages to phase 110 each time it receives a clockpulse signal. However, at large loads phase 120 may need to be triggeredbefore all of the current or energy in phase 110 is delivered to theload so the phaser circuit 156 sends a first trigger signal to phase 110and a second trigger signal to phase 120, as illustrated in more detailherein.

In some embodiments, relatively large error voltages (e.g., when Vout islower than Vref) indicate that more power is required at the load toraise Vout so the Voltage to Time Circuit decreases Tc so there is lesstime between triggering the phases 110, 120, and 130. Similarly, whenVout is near Vref there is a relatively smaller error voltage thatcorresponds to an increase in Tc and a longer time between sequentialphases, as described in more detail below.

In some configurations, the logic and control circuitry for one or moreof the Voltage to Time circuit 154, the phaser circuit 156, and thephases 110, 120, and 130 are physically combined with or intermixed withor near one another.

FIG. 4 is a waveform diagram illustrating wave forms for signals of thepower delivery control circuit 100 illustrated in FIG. 3. Now referringto FIG. 5, the waveforms for the power delivery control circuitillustrated in FIG. 3 are illustrated for a light load condition. Thefirst waveform is the inductor current (Lres Current) the secondwaveform is the voltage, the third waveform is the clock pulse and thefourth, fifth and sixth are the trigger signals for triggering phase 1,phase 2 and phase 3, respectively.

As shown in FIG. 4, for the given load condition, the Voltage to TimeCircuit has set the time between the starts of the phase charging timesas Tc. The phaser circuit triggers phase 110 to execute one cycle whichcauses an amount of charge or current or energy to be delivered to theinductor of the illustrated LC filter. After the time Tc has expired,the Voltage to Time Circuit 154 sends a second pulse to the phasercircuit 156. In this case the phaser circuit 156 triggers phase 120 toexecute one cycle which causes an amount of charge or current or energyto be delivered to the inductor of the illustrated LC filter. After thetime Tc has expired, the Voltage to Time Circuit 154 sends a third pulseto the phaser circuit 156. In this case the phaser circuit 156 triggersphase 130 to execute one cycle which causes an amount of charge orcurrent or energy to be delivered to the inductor of the illustrated LCfilter.

In some embodiments, the amount of charge or current or energy deliveredto the inductor by each of the phases 110, 120, and 130 may becontrolled. For example, the amount of charge or current or energy maybe controlled by design of the components of the power delivery controlcircuit 100, or may be controlled by controlling signals of the powerdelivery control circuit 100. For example, phaser circuit 156 may beconfigured to trigger phases 110, 120, and 130 by delivering pulses ofvariable widths, where the pulse width is controlled by a controller.Alternatively, each of phases 110, 120, and 130 may be configured todeliver a variable amount of charge or current or energy, where theamount is controlled by the controller. Other mechanisms of controllingthe charge or current or energy delivered to the inductor by each of thephases 110, 120, 130 may additionally or alternatively be used.

After the time Tc has expired, the Voltage to Time Circuit 154 sends afourth pulse to the phaser circuit 156. In this case the phaser circuit156 triggers phase 110 to execute one cycle which essentially sends acontrolled amount of energy to the connected inductor and out to theload. After the time Tc has expired, the Voltage to Time Circuit 154sends a fifth pulse to the phaser circuit 156. In this case the phasercircuit 156 decides to trigger phase 120 to execute one cycle whichessentially sends a controlled amount of energy to the connectedinductor and out to the load.

In some embodiments the power delivery control circuit 100 and/or phasercircuit 156 may have one or more of the following features:

Charge mode control, where the phaser circuit 156 is configured toarbitrate which phase to fire next. In some embodiments, the chargedelivered to the inductor during each cycle is controlled. In someembodiments, the current delivered to the output capacitor and loadstarts at zero and returns to zero in response to each pulse from theVoltage to Time Circuit 154.

At light loads, the time between a phase being fired again can be large.During an ‘off’ time between firing of phases, the phase is does notdeliver current to the inductor, and as such can be considered ‘off’ or‘shed’. In other words, phase shedding can be an automatic by product ofthis control scheme.

This auto phase shedding can also allow for portions of the phase to beturned off while the phase is shed. For example, a bias current to thephase may be turned down or off to reduce power consumption and heat.

In some embodiments some of the features of the power delivery controlcircuit 100 are:

Time based control loop architecture.

Each time a phase is ‘fired’ or ‘triggered’ it delivers a ‘quantity ofcharge’ to the output.

The control loop determines ‘Tc’, the time between successive phasetriggering. The smaller the ‘Tc’, the quicker next phase gets fired. Inother words, ‘Tc’ determines the rate at which ‘quantities’ of chargeget delivered to the output.

A Control Timer circuit can be configured to monitor the output voltageand the commanded/desired voltage (DAC voltage) to calculate the precise‘Tc’ required.

The Control Timer circuit can be digital, analog, or a combinationthereof.

The power delivery control circuit can utilize digital technology,analog technology or a combination of digital and analog technologies.More specifically, in some embodiments signals such as, but not limitedto, the error voltage and the output voltage can be analog signals ordigitized signals. The timer can be a digital programmed timer or ananalog timer that charges a capacitor. Similarly logic functions can beperformed with digital data or analog comparators. Any combination oftechnologies can be employed and this disclosure is in no way limited toa particular digital or analog technology to perform any particularfunction.

FIG. 5 is a waveform diagram illustrating wave forms for signals of thepower delivery control circuit 100 illustrated in FIG. 3. Now referringto FIG. 5, high load waveforms are illustrated for the power deliverycircuit shown in FIG. 3. As shown in FIG. 5, phase 110, phase 120 andphase 130 trigger pulses are much closer together such that a muchhigher average charge or current or energy is transferred to the loadthan for the low load case illustrated in FIG. 4. More specifically, thecharge or current or energy delivered by each phase overlaps in timewith the charge or current or energy delivered by each adjacent phase asshown by the Lres Current waveform.

FIG. 6 is a diagram illustrating an example embodiment of Tc and Verrdependence on load current. In alternative embodiments, the Tc and Verrdependence on load current has characteristics which are not illustratedin FIG. 6. For example, in some embodiments, the Tc and/or Verrdependence on load current is not linear. As shown in FIG. 6, as loadcurrent increases, Verr increases, and Tc reduces. In addition, asillustrated in FIG. 5, as load current increases, CLK pulses occur morefrequently. At high enough load current conditions the output of thephases overlap to provide an increased output current and can seamlesslytransition into continuous conduction mode (CCM) operation fordelivering even higher current. In some embodiments a CCM circuit suchas those disclosed in U.S. patent application Ser. No. 15/640,335 filedon Jun. 30, 2017 which is incorporated herein by reference in itsentirety for all purposes, can be used with the power delivery controlcircuit 100.

FIG. 7A is a schematic illustration of a power delivery engine 500. Inthis embodiment, power delivery engine 500 comprises a power regulatorcircuit. Now referring to FIG. 7A a non-limiting example schematic of apower delivery engine 500 that can be used for each phase of the powerdelivery control circuit 100 illustrated in FIG. 3 is shown. In thisexample the power delivery engine 500 includes a plurality ofsequentially coupled power switches M1, M2, M3, and M4.

FIG. 7B illustrates one example of the waveforms for the power deliveryengine 500 illustrated in FIG. 7A. These circuits and others aredescribed in more detail in U.S. Pat. No. 9,300,210 issued on Mar. 29,2016, which is incorporated by reference herein in its entirety for allpurposes.

Trace 805 illustrates a control voltage applied to first solid-stateswitch 130. In the particular embodiment depicted, switches are turnedon when approximately 1 volt is applied. At time t1 trace 805 is atapproximately 0 volts thus first solid-state switch 130 is off. Trace810 illustrates a control terminal voltage applied to second solid-stateswitch 140. At time t1 trace 810 is at approximately 1 volt thus secondsolid-state switch is on. Trace 815 illustrates a control terminalvoltage applied to third solid-state switch 150. At time t1 trace 815transitions to approximately 1 volt thus third solid-state switch 150transitions from off to on. Trace 820 illustrates a control terminalvoltage applied to fourth solid-state switch 160. At time t1 trace 820is at approximately 0 volts thus fourth solid-state switch is off.

Trace 825 illustrates a voltage at second junction 145. At time t1,capacitor 170 is shorted. Trace 830 illustrates current through inductor173. At time t1 inductor 173 is decoupled from the remainder of switchedregulation circuit 125 thus the current in inductor 173 is zero. Trace835 illustrates a comparator output corresponding to a zero currentcondition in inductor 173, as discussed in more detail below. Trace 840illustrates the voltage across capacitor 170. At time t1 capacitor 170is shorted causing the voltage across capacitor 173 to decrease as thecapacitor is discharged.

Now referring to FIG. 8 a non-limiting example schematic of a powerdelivery engine 600 that can be used in each phase of the circuit 100illustrated in FIG. 3 is shown. In this example the power deliveryengine 600 includes a two coupled power switches 610 and 620 arranged ina synchronous buck converter configuration, as known in the art. Otherembodiments can have different power delivery engines including, but notlimited to full bridge, buck, boost, buck boost, and other types ofpower control circuits known by one of skill in the art.

Now referring to FIG. 9 a transient performance boost circuit 700 isshown that can be used instead of control circuit 150 in the powerdelivery control circuit 100 illustrated in FIG. 3. In some embodimentsthe transient performance boost circuit 700 can include one or more ofthe following features:

For fast transient response, the error voltage generation 710 canincorporate multiple enhancement schemes.

One such scheme temporarily increases the GM of an error amplifier inthe Error Voltage generation 710. The increase in GM helps the controlvoltage “Ve” quickly ramp up or down in response to the error voltageseen at the input (Vout-Vdac).

Another scheme employs a feed forward signal (Ie) supplied from theerror amplifier and delivered to the timer such that during a transientan error current bypasses the compensation network 720 and quicklychanges the timer circuit 730 output (Increases or decreases thefrequency of phase firing).

Another scheme employs a fixed offset in the timer circuit 730.Detecting a transient, the offset can be either increased or decreasedinstantly such that the frequency change is instant giving a rapidresponse.

Another scheme employs multiple bands of frequency of operation. Timercircuit 730 is configured to generate a range of output frequenciescorresponding with the functional range of control voltage (Ve). In amultiple frequency band scheme, timer circuit 730 is programmable so asto generate a different range of output frequencies for each frequencyband. Accordingly, when using a multiple frequency band scheme, thefrequency generated by timer circuit 730 is determined based on both thecontrol voltage (Ve) and the programmed frequency band. Control of theoutput frequency is achieved through a combination of response to errorvoltage for fine control and frequency band selection for coarsecontrol.

During transient conditions, bands can be hopped (band hopping) up ordown to quickly achieve the desired operating frequency. For example, inresponse to the control voltage (Ve) saturating, by being driven outsideits functional range, timer circuit 730 may be programmed by acontroller to operate in an appropriate adjacent higher or lowerfrequency band. Alternatively or additionally, timer circuit 730 may beprogrammed by a controller to operate in a higher or lower frequencyband in response to the control voltage (Ve) increasing or decreasing ata rate greater than a threshold.

In some embodiments bump type sequencing of each phase can be used whena plurality of serially coupled power devices are used, for example, asdescribed in application Ser. No. 15/640,335 filed on Jun. 30, 2017 andU.S. Pat. No. 9,300,210 issued on Mar. 29, 2016, which are incorporatedherein in their entirety by reference. The bump type sequencing caninclude one or more of the following features:

Each phase can deliver two types of “Bumps” or charge deliverysequences, named VDD bump and GND bump, a VDD bump caused by currentsourced from or sunk by the VDD power supply being sunk by or sourcedfrom the connected inductor, and a GND bump caused by current sourcedfrom or sunk by the connected inductor being sunk by or sourced from theGND power supply.

The phases may collectively deliver these two bumps strictly inalternate sequence. VDD=>GND=>VDD=>GND and so on. For example, this mayoccur by each of the phases delivering alternating bumps(VDD=>GND=>VDD=>GND and so on.)

In some embodiments, an alternative bump sequence may be preferred.

For example, a VDD bump may excite the supply network more than a GNDbump. Therefore, the phases in a multiphase system may collectivelydeliver more GND bumps than VDD bumps.

In some systems, a frequency of the VDD bumps and/or the GND bumps maybe controlled through selection of a VDD/GND bump sequence.

In some embodiments the phase firings can be sequenced to achieve anarbitrary bump sequence. For instance a 3 phase system can deliverVDD=>VDD=>VDD=>GND=>GND=>GND sequence. Or it can also deliverVDD=>GND=>VDD=>GND=>VDD=>GND sequence. The bump sequence used can affectthe frequency at which the input network is excited.

Depending on the input impedance network, an optimal choice of bumpsequence can be programed to achieve the optimal supply noisecharacteristics.

The power supply switching frequency can be kept away from the inputnetwork resonant frequency (or where the input impedance is large).

This feature can also help during transient response at least becausethe current load on the power supplies are distributed in time. As aresult, the bypass capacitance and low power bus impedance aresufficient to prevent unacceptable power supply transients. Therefore,the voltage difference between the positive and negative power suppliesremains substantially fixed.

In some embodiments, the bump sequence of each of the phases iscontrolled using methods discussed in described in application Ser. No.15/640,335 filed on Jun. 30, 2017 and U.S. Pat. No. 9,300,210 issued onMar. 29, 2016, referenced above. To coordinate a collective bumpsequence collectively generated by the multiple phases, a controllerreceives or determines a target collective bump sequence, and determinesa bump sequence for each of the phases. The controller provides signalsfor each of the individual phases so as to cause each of the individualphases to operate with the bump sequence determined therefor by thecontroller. Accordingly, each of the individual phases operate with thebump sequence determined therefor, and the collective bump sequencegenerated by the multiple phases correspondence with the targetcollective bump sequence.

Now referring to FIG. 10, a comparator mode control circuit 800 can beused instead of control circuit 150 in the power delivery controlcircuit 100 illustrated in FIG. 3. The comparator mode control circuit800 can combine the output of the timer 810 with the output of acomparator 820 so the decision to trigger the next phase includes thefollowing conditions: 1) has the timer expired? and 2) is the outputvoltage below a threshold voltage? Both of these conditions must be truefor the next phase to trigger. This feature can be particularly usefulfor semiconductor processes where a timer with a wide time range isdifficult to make so at light loads the comparator can be relied uponsuch that the next phase will only be triggered if the output voltagegoes below a threshold voltage. In some embodiments the comparator modecontrol can have one or more of the following features:

Comparator mode control circuit 800 may be used in addition to one ormore other control schemes. For example, comparative mode controlcircuit 800 and control circuit 150 may both be used. Which controlcircuit is active to be determined, for example, based on loadconditions. For example, comparator more control mode control circuit800 may be used when the load is less than a threshold.

Band hopping along with a comparator can be used to provide a fasttransient response.

The comparator mode control is also useful during Start up, DynamicVoltage Scaling (DVS) Up and DVS Down. In comparator mode overshoot andundershoot is minimized based on the state of the comparator output.

In some embodiments, the comparator 820 is hysteretic.

In some embodiments a comparator control circuit can be included as aportion of the phaser circuit. More specifically, a comparator controlcircuit can use Vout and the clock signal to only allow the phaser toexecute a phase if Vout is below a predetermined voltage and a clocksignal is received from the Voltage to Time circuit. This feature canprotect against the phaser sending trigger signals to one or more phasesif Vout is above the predetermined voltage but due to transients orbandwidth limitations of the control circuit one or more clock signalsare sent. Because Vout is above the predetermined voltage no phases willbe triggered.

In some embodiments any logical combination of Vout and the timer outputcan be used as an input to the phaser. In various embodiments thecomparator control circuit can be implemented via analog circuitry,digital circuitry or a combination thereof. In one example, the outputvoltage can be digitized, the timer can be digital and a digitalprocessor can be used to make a logical decision whether or not totransmit a pulse to the phaser.

Now referring to FIGS. 11 and 12, an embodiment of a voltage to timecircuit 900 is illustrated. Voltage to time circuit 900 can be used withthe power delivery control circuit shown in FIG. 3. In some embodimentsthe voltage to time circuit 900 can have one or more of the followingfeatures:

The trip voltage can be dynamically changed to get faster responseduring a transient. For example, during a loading transient trip voltagecan be decrease.

For DVS up transition, trip voltage can be decreased while for a DVSdown transition, trip voltage can be increased.

The capacitance of capacitor C can be changed to increase or decreasethe clock output frequency

FIGS. 13 and 14 illustrate an embodiment of a voltage to time circuit.Now referring to FIGS. 13 and 14, an embodiment of a voltage to timecircuit 1100 is illustrated. Voltage to time circuit 1100 can be usedwith the power delivery circuit shown in FIG. 3. In some embodiments thevoltage to time circuit can have one or more of the following features:

Timer current can be programmable to give control on the clock frequencyrange.

Timer current can be dynamically changed to improve transient response,for example, during loading transient, the timer current can beincreased to generate faster clock frequency.

During DVS up, timer current can be increased.

The capacitance of capacitor C can be changed to increase or decreasethe clock output frequency.

FIGS. 15 and 16 illustrate an embodiment of inductor shorting circuitLshort. In this embodiment inductor shorting circuit Lshort is a switch.Now referring to FIGS. 15 and 16, in some embodiments inductor shortingcan be used to improve linearity of the power delivery circuit shown inFIG. 3. Inductor shorting can include one or more of the followingfeatures:

During discontinuous current mode (DCM) operation, the phase circuit ofFIG. 15 presents a high impedance to the switching node Vx. As a result,the voltage at switching node Vx and the current through the inductorring based on the capacitances, resistances, and inductances of thecircuit according to principles understood by those of skill in the art.The ringing may not be desired in some embodiments since it leaves thestarting current in the inductor at the beginning of the next cycle,when switching node Vx is again driven by the phase, in an uncontrolledstate. The uncontrolled state is at least partially influenced byprevious data, such that the uncontrolled state causes non-linearity.

An inductor shorting circuit Lshort can be used such that while theinductor is not driven by the phase, the switching node Vx and theoutput Vout are shorted.

The shorting causes the inductor current to be equal or substantiallyequal to zero. This allows for the next cycle of that phase, whenswitching node Vx is again driven by the phase, to begin with acontrolled and/or consistent zero or substantially zero or near zerocurrent instead of starting in the uncontrolled state, which may beeither a positive or a negative current.

To short the inductor, inductor shorting circuit Lshort becomesconductive while the phase connected thereto is in a high impedancestate (for example, as is common to multiple parallel coupled FETS andBuck architectures). Inductor shorting circuit Lshort may remain on fora small predetermined amount of time or may stay on until just beforethe phase is fired again.

The inductor shorting circuit Lshort shorts the output inductor andprovides a low impedance path for charging the parasitic capacitance atswitching node Vx to the voltage at node Vout.

Because Lout and Cout form an LC oscillator, without the inductorshorting circuit Lshort, the parasitic node may ring undesirably.

In another embodiment, the inductor shorting circuit Lshort canelectrically short the switching node Vx to another voltage source(Supply for example) for a brief amount of time to charge the switchingnode Vx to the supply voltage, after which the inductor shorting circuitLshort may be opened. By charging the switching Vx node to the supplyvoltage, the ringing may be greatly reduced and/or may be controlledsuch that the next cycle of the phase, when switching node Vx is againdriven by the phase, begins with a consistent current instead ofstarting in the uncontrolled state.

Now referring to FIG. 9 a transient performance boost circuit 700 isshown that can be used instead of control circuit 150 in the powerdelivery control circuit 100 illustrated in FIG. 1. In some embodimentsthe transient performance boost circuit 700 can include one or more ofthe following features:

FIG. 17 is a schematic illustration of a control timer circuit 1500. Nowreferring to FIG. 17, a control timer 1500 is shown that can be usedinstead of control circuit 150 in the power delivery control circuit 100illustrated in FIG. 3. In this embodiment, a compensation network 1520is coupled between the Error Amplifier 1510 and the Voltage to Timecircuit 1530. In various embodiments the compensation network 1520 isused to improve the stability of the Verr signal to make the feedbackloop more stable and reliable. In one example embodiment thecompensation network 1520 can include a capacitor to ground as shown. Infurther embodiments, illustrated in FIGS. 18, 19 and 20, thecompensation network can include a Type 3, a Type 2 or a Type 1compensation circuit, respectively, as known by those of skill in theart. Other compensation networks can also be used and this disclosure isin no way limited to the disclosed example compensation networks. Forexample, as illustrated in FIG. 17, another scheme employs a feedforward signal (Ie) from the error amplifier 1510 to the timer 1530 suchthat during a transient an error current bypasses the compensationnetwork and quickly changes the timer output (e.g., increases ordecreases the frequency of phase firing).

In some embodiments one or more telemetry features can be implementedfor the power delivery control circuitry 100 illustrated in FIG. 3. Forexample, in one embodiment telemetry circuitry can be configured torecord digital or analog data from the power delivery and controlcircuitry that corresponds to the current output, voltage output orother characteristic of the power delivery and control circuitry. Insome embodiments the power delivery and control circuitry 100 can beused in conjunction with an integrated circuit that includes a processorwherein the processor can be commanded to record the telemetry data andstore the telemetry data in an associated memory. In various embodimentsthe telemetry data can be recorded only when commanded, or in otherembodiments it can be recorded continually, for example when used inconjunction with a FIFO memory.

In some embodiments the telemetry circuitry can acquire data associatedwith the current output of the power delivery and control circuitry 100by recording data representing the Verr signal generated by the ErrorAmplifier 152 since the Verr signal can be correlated to the outputcurrent. In other embodiments the telemetry circuitry can acquire datacorresponding to the frequency of the clock pulses sent by the Voltageto Time (Tc) circuit 154 which can also be correlated with the outputcurrent. The accuracy of the telemetry data and how precisely it iscorrelated to the actual current delivered by the power delivery andcontrol circuitry can be affected by how well controlled and/or knownthe characteristics of the components of the power delivery and controlcircuitry are. For example, the specific values of the capacitors,inductors and resistors can affect the accuracy of the data, thereforeto improve the accuracy the tolerance on such components can be eitherhighly controlled and/or the components can be characterized and thesystem can be trimmed to compensate for the characteristics, therebyimproving the accuracy.

For simplicity, various peripheral electrical components are not shownin the figures.

Regulator with Continuous Current

In some embodiments power delivery control circuit 100 (see FIG. 3) canbe configured to provide continuous current and/or an increase incurrent to the load by continuously maintaining the current in theinductor of at least one of the phases 110, 120, 130 above zero, asdescribed in more detail below.

Now referring simultaneously to FIGS. 7A, and 19-26 an example switchingsequence and timing diagram for an embodiment of switched regulationcircuit 125 (see FIG. 7A) with continuous and/or increased current isillustrated. More specifically, FIG. 7A illustrates a simplifiedschematic of switched regulation circuit 125; FIG. 21 illustrates anexample switching sequence 1900 having sequential steps 1905 through1940 for the four switches in switched regulation circuit 125; FIG. 22illustrates an example timing diagram showing the control signalsdelivered to each of the four solid-state switches as well as thecurrent within inductor 173 (IL), and the voltage at second junction 145(V145) during switching sequence 1900; and FIGS. 23-28 illustratesimplified circuit schematics of each of the six different solid-stateswitch configurations described in switching sequence 1900. In FIGS.23-28 solid-state switches that are in an on state are depicted withsolid lines and solid-state switches that are in an off state aredepicted with dashed lines. The switching sequence illustrated in FIG.21 is for example only and other switching sequences, timings andconfigurations are within the scope of this disclosure.

Now referring to FIG. 21, switching sequence 1900 having sequentialsteps 1905 through 1940 is illustrated. In step 1905, first, second andthird solid-state switches M1, M2 and M3, respectively, are controlledto be in an on state and fourth solid-state switch M4 is controlled tobe in an off state. A simplified schematic of switched regulationcircuit 125 in step 1905 is illustrated in FIG. 23. Step 1905 is a firstinductor prefluxing state where current in inductor 173 (see FIG. 7A) isincreased at a substantially linear rate by the application of the inputvoltage at first terminal 120 (Vin) across the inductor, at a timebefore capacitor 170 is charged.

Example currents and voltages within switched regulation circuit 125 forstep 1905 are illustrated in timing diagram 2000 (see FIG. 22). Thelogic levels for solid-state switch control signals M1, M2, M3, M4 areindicated by traces 2005, 2010, 2015 and 2020, respectively. A highlogic level (sometimes noted as 1) indicates the switch (or compositeswitch) is in an on state, and a low logic level (sometimes noted as 0)indicates the switch is in an off state.

Timing diagram 2000 illustrates that first step 1905 occurs at time t1.At time t1, trace 2005 shows that a high logic level control signal isapplied to first solid-state switch 130, placing it in an on state.Trace 2010 illustrates that at time t1 a high logic level control signalis applied to second solid-state switch 140, placing it in an on state.Trace 2015 illustrates that at time t1 a high logic level control signalis applied to third solid-state switch 150, placing it in an on state.Trace 2020 illustrates that at time t1 a low logic level control signalis applied fourth solid-state switch 160, placing it in an off state.

Continuing to refer to timing diagram 2000, at t1 a voltage at secondjunction 145 (see FIG. 7A) is illustrated by trace 2025 and issubstantially equivalent to the Vin voltage (minus a relatively smallvoltage drop across M1 and M2) at first node 120. Current in inductor170 (IL trace 2030) increases rapidly, corresponding to the appliedvoltage and the characteristics of inductor 173. For some embodiments,the voltage at node 176 (see FIG. 7A) may change a relatively smallamount compared with the voltage across the inductor and thus thecurrent may increase substantially linear at a rate approximated by(Vin−Vout)/L where Vout is the voltage at node 176. The current ininductor 173 continues to increase while in this switch state, theduration of which may be controlled by a timer, shown in step 1910 as adelay.

In some embodiments the timer in step 1910 can be fixed while in otherembodiments it can be a variable timer. In one example the variabletimer can use a lookup table to adjust according to different loadconditions and demands on switched regulation circuit 125. Morespecifically, in some embodiments the timer can be set proportional to a“duty factor” (e.g., Vout/Vin) such that a longer amount of time isselected when a higher Vout is required. In further embodiments thetimer in step 1910 can be controlled by a feedback loop based on one ormore characteristics of switched regulation circuit 125. In someembodiments the timer may be adjusted by the feedback loop to energizeinductor 173 with an appropriate amount of current so that the inductorcurrent resonates to a predetermined current when the first resonatingstate is engaged (discussed in the next step 1915). In furtherembodiments the timer can use a comparator that compares the current inthe inductor to a programmable current threshold.

In other embodiments, the timer can be made utilizing a current on acapacitor wherein the current starts charging at the beginning of thepreflux cycle and may be compared to a predetermined voltage. When thevoltage on the capacitor reaches a specified voltage the timer indicatesthat the preflux cycle should end. In other embodiments the timerfunction can be performed utilizing logic gates.

In some embodiments, instead of a timer for setting the amount ofpreflux, the current in the inductor can be monitored during preflux(e.g., step 1905) and the preflux cycle can be set to end when thecurrent reaches a specified level. That specified current level can beadjusted on a cycle by cycle basis to optimize performance. Other timertechniques and timer architectures can be used and are within the scopeof this disclosure.

Now referring to FIG. 21, after the delay in step 1910, the controlleradvances to step 1915 where first and third solid-state switches M1 andM3 remain on while the second solid-state switch M2 is turned off andthe fourth solid-state switch M4 remains off. Thus, first and thirdsolid-state switches, M1, M3, respectively, are on while second andfourth solid-state switches M2, M4, respectively, are off. A simplifiedschematic of switched regulation circuit 125 in step 1915 is illustratedin FIG. 24. This state couples capacitor 170 in series with inductor 173and the voltage at first terminal 120 (Vin) is applied directly tosecond junction 145. Capacitor 170 now begins to charge. Capacitor 170charges with a time constant set by capacitor 170 and inductor 173values. Further, as capacitor 170 begins to charge, current flow ininductor 173 continues to increase as the voltage between secondjunction 145 and the output is positive. Because of the prefluxingoperation in step 1905, the current that was already flowing in inductor173 continues to increase, as discussed in more detail below.

Step 1915 is illustrated in timing diagram 2000 (see FIG. 22) at timet2. Now referring simultaneously to FIGS. 7A and 20, at time t2, secondsolid-state switch 140 (i.e., trace 2010) turns off. The voltage atsecond junction 145 (i.e., trace 2025) begins to decrease. Current ininductor 173 (trace 2030) continues to build as capacitor 170 charges.Voltage in capacitor 170 increases towards Vin. As capacitor 170 becomescharged the current in inductor 173 (trace 2030) peaks, then begins todecrease when the voltage at node 145 equals the voltage at node 176 andcontinues to decrease towards time t3. Thus, in step 1915, capacitor 170charges, causing a current to flow in inductor 173, and increasing thevoltage at output node 176 (Vout). When capacitor 170 is fully chargedto the voltage at (Vin) 120, the controller proceeds to step 1920 (seeFIG. 21) which is a first “soft braking” configuration that can be usedto transition the current remaining in inductor 173. Soft braking canenable a higher current per phase and/or a smaller capacitor 170 perphase as compared to the methodologies discussed above and as explainedin more detail below.

In the first soft braking configuration (step 1920) first, third andfourth solid-state switches M1, M3 and M4, respectively, are on whilesecond solid-state switch M2 is turned off. A simplified schematic ofswitched regulation circuit 125 in step 1920 is illustrated in FIG. 25.In this state inductor 173 is coupled to Vin (node 120) throughcapacitor 170 and also to ground 165 through third and fourthsolid-state switches, M3 and M4, respectively, allowing the residualcurrent in the inductor to continue to decrease down to a non-zerominimum current (Imin).

Step 1920 is illustrated in timing diagram 2000 (see FIG. 22) at timet3. Now referring simultaneously to FIGS. 7A and 20, at time t3, fourthsolid-state switch 160 (i.e., trace 2020) turns on adding a path betweeninductor 173 and ground 165. The voltage at second junction 145 (i.e.,trace 2025) remains at the ground potential and current in inductor 173(trace 2030) continues to decrease as the inductor dissipates its storedenergy. Current in inductor 173 continues to decrease to a predeterminedminimum current (Imin) that is non-zero in this particular embodiment.In some embodiments the minimum current (Imin) can be between 10milliamps and 50 amperes, while in other embodiments it can be between100 milliamps and 1 ampere and in another embodiment it can be between200 milliamps and 400 milliamps. The controller then proceeds to step1925 (see FIG. 21) that is a second prefluxing state that can be used toincrease current flowing through inductor 173.

Now referring to FIG. 21, in step 1925 first fourth solid-stateswitches, M1 and M4 remain on, second solid-state switch M2 turns on,and third solid-state switches M3 remains off. A simplified schematic ofswitched regulation circuit 125 in step 1925 is illustrated in FIG. 26.This is the second inductor prefluxing stage where current in inductor173 is increased at a substantially linear rate by applying voltage atfirst output terminal 120 (Vin) to the inductor. In this state thevoltage at first terminal 120 (Vin) is applied directly across inductor173.

Now referring to timing diagram 2000, the second prefluxing state (step1925) is shown at t4. The voltage at second junction 145 rapidlyincreases to the Vin voltage at first node 120 shown by trace 2025.Current in inductor 170 (trace 2030) increases rapidly, corresponding tothe applied voltage and the characteristics of inductor 173. In someembodiments the rate of current increase can be substantially similar tothe rate of current increase in the time between t1 and t2. The currentin inductor 173 continues to increase until the switch state is changed,which in one embodiment, may be controlled by a delay shown in step 1930that can be controlled by a timer, as discussed above.

Now referring to FIG. 21, in step 1935 fourth solid-state switch M4remains on and second solid-state switch M2 is turned on while first andthird solid-state switches M1, M3, respectively, remain off. Asimplified schematic of switched regulation circuit 125 in step 1935 isillustrated in FIG. 27. Capacitor 170 is connected between inductor 173and ground 165, allowing the charge stored in the capacitor to dischargethrough the inductor to load 115 (see FIG. 1). As capacitor 170 beginsto discharge (with a time constant set by capacitor 170 and inductor173), current in inductor 173 increases. This condition is illustratedin timing diagram 2000 in FIG. 22 at time t5 showing the voltage atsecond junction 145 (i.e., trace 2025) at a voltage that is close to thevoltage at Vin (120) as it is connected to fully charged capacitor 170.As capacitor 170 resonates with inductor 173, it discharges causingcurrent to increase in inductor 173 (i.e., trace 2030). The increase incurrent causes the voltage at Vout (node 176) to increase. As the chargestored in capacitor 170 is reduced, current in inductor 173 peaks(Ipeak), then begins to decrease (trace 2030).

The controller then proceeds to step 1940 (see FIG. 21) which is asecond “soft braking” configuration that can be used to transition theremaining current in inductor 173. Soft braking can enable a highercurrent per phase and/or a smaller capacitor 170 per phase as discussedabove.

More specifically, in step 1940 second, third and fourth solid-stateswitches M2, M3 and M4, respectively, are on while first solid-stateswitch M1 is turned off. A simplified schematic of switched regulationcircuit 125 in step 1935 is illustrated in FIG. 28. In this stateinductor 173 is coupled to ground 165 through third and fourthsolid-state switches, M3 and M4, respectively, allowing the residualcurrent in the inductor to continue to decrease down to a non-zerominimum current (Imin).

Step 1940 is illustrated in timing diagram 2000 (see FIG. 22) at timet6. Now referring simultaneously to FIGS. 7A and 20, at time t6, thirdsolid-state switch 150 (i.e., trace 2015) turns on adding a path betweeninductor 173 and ground 165. The voltage at second junction 145 (i.e.,trace 2025) remains at the ground potential and current in inductor 173(trace 2030) continues to decrease as the inductor dissipates its storedenergy. Current in inductor 173 continues to decrease to a predeterminedminimum current (Imin) that is non-zero in this particular embodiment.The controller then returns to step 1905 (see FIG. 21) which is thefirst prefluxing state that can be used to increase current flowingthrough inductor 173.

Timing diagram 2000 in FIG. 22 is for illustration only and is oneexample of the function of circuit 125 (see FIG. 7A) operating with anon-zero inductor current. Other switching algorithms, control functionsand features can be implemented without departing from this disclosure.To control the duration of any of steps 1905-1940 illustrated in FIG. 21any type of timer or control circuit can be used, including thosedisclosed herein. For example, in some embodiments a comparator can beused to compare output voltage (Vout) to a commanded voltage. If theoutput voltage is too low the controller can shorten the soft brakeduration and start the next preflux step early, leading to a higheroutput voltage and higher average output current delivered to load (seeFIG. 7A). This control algorithm can also provide a relatively fastresponse time to loads having high transient voltage requirements. Infurther embodiments a multi-phase architecture can be employed wheremultiple switched regulation circuits 125 (see FIG. 2) are used togetherto provide power to load 115.

In further embodiments alternative switching sequences 1900 can be usedto provide additional features and functions of switched regulationcircuit 125 (see FIG. 7A). For example, wait states can be added afterfirst and second soft brake steps (steps 1920 and 1940, respectively) toprovide light load performance. More specifically, when load 115 (seeFIG. 7A) requires a reduced amount of current and/or voltage, afterfirst softbrake (step 1920) a wait state can be commanded where firstand fourth solid-state switches, M1 and M4, respectively are on andsecond and third solid-state switches, M2 and M3 are off. Thisessentially halts current flow through circuit 125 to load 115 (see FIG.7A) until the subsequent preflux step 1925 is commanded. Similarly,after second soft brake (step 1940) a second wait state can be commandedwhere second and third solid-state switches, M2 and M3, respectively,are on and first and fourth solid-state switches, M1 and M4,respectively, are off. This state essentially halts current flow throughcircuit 125 until the subsequent preflux step 1905 is commanded.

In some embodiments a comparator control circuit can be included as aportion of the phaser circuit. More specifically, a comparator controlcircuit can use Vout and the clock signal to only allow the phaser toexecute a phase if Vout is below a predetermined voltage and a clocksignal is received from the Voltage to Time circuit. This feature canprotect against the phaser sending trigger signals to one or more phasesif Vout is above the predetermined voltage but due to transients orbandwidth limitations of the control circuit one or more clock signalsare sent. Because Vout is above the predetermined voltage no phases willbe triggered.

In the foregoing specification, embodiments of the disclosure have beendescribed with reference to numerous specific details that can vary fromimplementation to implementation. The specification and drawings are,accordingly, to be regarded in an illustrative rather than a restrictivesense. The sole and exclusive indicator of the scope of the disclosure,and what is intended by the applicants to be the scope of thedisclosure, is the literal and equivalent scope of the set of claimsthat issue from this application, in the specific form in which suchclaims issue, including any subsequent correction. The specific detailsof particular embodiments can be combined in any suitable manner withoutdeparting from the spirit and scope of embodiments of the disclosure.

Additionally, spatially relative terms, such as “bottom or “top” and thelike can be used to describe an element and/or feature's relationship toanother element(s) and/or feature(s) as, for example, illustrated in thefigures. It will be understood that the spatially relative terms areintended to encompass different orientations of the device in use and/oroperation in addition to the orientation depicted in the figures. Forexample, if the device in the figures is turned over, elements describedas a “bottom” surface can then be oriented “above” other elements orfeatures. The device can be otherwise oriented (e.g., rotated 90 degreesor at other orientations) and the spatially relative descriptors usedherein interpreted accordingly.

Various details are set forth herein as they relate to certainembodiments. However, the invention can also be implemented in wayswhich are different from those described herein. Modifications can bemade to the discussed embodiments by those skilled in the art withoutdeparting from the invention. Therefore, the invention is not limited toparticular embodiments disclosed herein.

Though the present invention is disclosed by way of specific embodimentsas described above, those embodiments are not intended to limit thepresent invention. Based on the methods and the technical aspectsdisclosed herein, variations and changes may be made to the presentedembodiments by those of skill in the art without departing from thespirit and the scope of the present invention.

What is claimed is:
 1. A system comprising: a substrate; a first chip onthe substrate, wherein a load circuit is integrated on the first chip;and a second chip on the substrate, wherein a power delivery circuit isconfigured to deliver current to the load circuit according to aregulated voltage at a node, wherein the power delivery circuitcomprises: a first circuit configured to generate an error signal basedat least in part on the regulated voltage, and a voltage generatorcomprising power switches configured to modify the regulated voltageaccording to the error signal, wherein the first circuit of the powerdelivery circuit is integrated on the first chip, and wherein at least aportion of the power switches of the power delivery circuit areintegrated on the second chip.
 2. The system of claim 1, wherein theerror signal represents a difference between the regulated voltage atthe node and a reference voltage.
 3. The system of claim 2, wherein theerror signal comprises a series of pulses, and wherein a frequency ofthe series of pulses is based on the difference.
 4. The system of claim1, wherein the error signal is an analog voltage.
 5. The system of claim1, wherein the error signal is a digital value.
 6. The system of claim1, wherein the error signal represents a difference between theregulated voltage at the node and a reference voltage multiplied byagain factor.
 7. The system of claim 1, wherein the first circuitcomprises an analog-to-digital converter configured to generate theerror signal.
 8. The system of claim 1, wherein the power deliverycircuit comprises a capacitor connected to the load circuit, wherein thecapacitor is integrated on the first chip.
 9. The system of claim 1,wherein the power delivery circuit comprises one or more inductorsconnected to the load circuit, wherein the inductors are formed on thesubstrate separate from the first and second chips.
 10. The system ofclaim 1, wherein all of the power switches of the power delivery circuitare integrated on the second chip.
 11. The system of claim 1, furthercomprising a reference voltage generator configured to generate areference voltage, wherein the error signal represents a differencebetween the regulated voltage at the node and the reference voltage andwherein the reference voltage generators integrated on the first chip.12. The system of claim 1, wherein the power delivery circuit comprises:a capacitor connected to the load circuit; and one or more inductorsconnected to the load circuit, wherein the power switches, thecapacitor, and the one or more inductors collectively form a voltageregulator.
 13. The system of claim 12, wherein the voltage regulator ismultiphase.
 14. A method of forming a system, the method comprising:attaching a first chip to a substrate, wherein a load circuit isintegrated on the first chip; and attaching a second chip to thesubstrate, wherein a power delivery circuit is configured to delivercurrent to the load circuit according to a regulated voltage at a node,wherein the power delivery circuit comprises: a first circuit configuredto generate an error signal based at least in part on the regulatedvoltage, and a voltage generator comprising power switches configured tomodify the regulated voltage according to the error signal, wherein thefirst circuit of the power delivery circuit is integrated on the firstchip, and wherein at least a portion of the power switches of the powerdelivery circuit are integrated on the second chip.
 15. The method ofclaim 14, wherein the error signal represents a difference between theregulated voltage at the node and a reference voltage.
 16. The method ofclaim 15, wherein the error signal comprises a series of pulses, andwherein a frequency of the series of pulses is based on the difference.17. The method of claim 14, wherein the error signal is an analogvoltage.
 18. The method of claim 14, wherein the error signal is adigital value.
 19. The method of claim 14, wherein the error signalrepresents a difference between the regulated voltage at the node and areference voltage multiplied by again factor.
 20. The method of claim14, wherein the first circuit comprises an analog-to-digital converterconfigured to generate the error signal.
 21. The method of claim 14,wherein the power delivery circuit comprises a capacitor connected tothe load circuit, wherein the capacitor is integrated on the first chip.22. The method of claim 14, wherein the power delivery circuit comprisesone or more inductors connected to the load circuit, wherein theinductors are formed on the substrate separate from the first and secondchips.
 23. The method of claim 14, wherein all of the power switches ofthe power delivery circuit are integrated on the second chip.
 24. Themethod of claim 14, further comprising a reference voltage generatorconfigured to generate a reference voltage, wherein the error signalrepresents a difference between the regulated voltage at the node andthe reference voltage and wherein the reference voltage generatorsintegrated on the first chip.
 25. The method of claim 14, wherein thepower delivery circuit comprises: a capacitor connected to the loadcircuit; and one or more inductors connected to the load circuit,wherein the power switches, the capacitor, and the one or more inductorscollectively form a voltage regulator.
 26. The method of claim 25,wherein the voltage regulator is multiphase.